This invention relates to a ballast for lighting devices, and in particular, to an electronic ballast for discharge lamps.
The most efficient electrical lighting sources that are commonly available are gaseous, low pressure and high-pressure discharge lamps. Examples of these include high intensity discharge (HID) lamps. These types of lamps typically utilize a gas sealed within a tube, which gives off light when excited with an electrical signal.
Electronic ballasts typically include switching transistors and utilize high switching frequencies to convert energy into an appropriate form to drive the lamps. For HID lamps, it is common to drive the lamps with a low frequency (for example 100 Hz) square wave of current. Electronic ballasts of this type typically switch transistors off and on utilizing a duty cycle selected to adjust the power delivered to the lamp. More specifically, the power that the ballast delivers is typically determined from the duty cycle of switching transistors, wherein a greater duty cycle implies a higher amplitude driving current, which results in more power and thus a brighter light output.
FIG. 1 shows a prior art technology described U.S. Pat. No. 5,917,290, issued to Shen, the applicant herein. In accordance with FIG. 1, an output light source Vout is driven by an input voltage Vin through a ballast control circuit as shown. In such systems, the switching cycle of the bridge transistors M3 and M4 is synchronized with the input mains voltage. Switches M1, M2, M5 and M6 are operated at high frequency in order to shape the current drawn from the mains and to deliver a square wave of current to the lamp that is synchronized to the mains.
In operation, the switches M1 through M6 operate in conjunction with each other, with M1, M2, M5 and M6 operated in a high frequency pulse width modulation mode (PWM) to drive the output light source Vout In such an arrangement, the current through L1 is regulated by the switching of transistors M1 and M2 in a manner such that a sinusoidal wave current is present in L1. The sinusoidal wave effectively follows the input voltage Vin, scaled by a prescribed factor. Similarly, the current through L2 is regulated by the switching of transistors M5 and M6 in a manner such that a square wave of current is present in L2.
In order to control the power drawn from the input, the current through inductor L1 must be monitored, and the duty cycle utilized to switch transistors M1 and M2 appropriately adjusted through a feed-back loop in order to provide the proper amplitude.
FIG. 1A shows the representation of the current flowing through inductor L1 of FIG. 1, superimposed upon the switching state of the transistor M2. As FIG. 1A shows, the current within inductor L1 is substantially constant for any switching cycle, a switching cycle referring to the high frequency switching cycles of M1 and M2.
Similarly, in order to control the power delivered to the output, the current through inductor L2 must be monitored, and the duty cycle utilized to switch transistors M5 and M6 appropriately adjusted through a feedback loop in order to provide the proper amplitude.
Although the arrangement of the ""290 patent solves many of the prior art problems, there are still two problems that the ""290 arrangement does not solve. First, as previously indicated, the amplitude of the current flowing through L1, and thus of the power drawn from the input, is controlled by rapidly switching the transistors off and on. The particular sequence of switching these transistors off and on as described in the ""290 patent and other prior art systems results in significant power loss through the ballast. A similar situation exists with inductor L2 and switches M5 and M6. Accordingly, the arrangement is less efficient than desired.
Second, the technique utilized for matching the current through inductor L1 to a specified AC voltage is a feedback loop. More specifically, the current through L1 is monitored, fed back to an error amplifier, and the output of such error amplifier is utilized to adjust the duty cycle of the transistors in order to increase or decrease the current through L1 to the desired value. A similar feedback loop arrangement must be utilized for the current in L2. The constant monitoring of the currents through L1 and L2 and the use of the feedback loops requires additional components (not shown in FIG. 1) which add to the cost and complexity of the ballast circuit.
In view of the above, there exists a need in the art for a more cost effective manner of controlling a ballast to drive a lighting device.
The above and other problems of the prior art are overcome and a technical advance achieved in accordance with the present invention. An electronic ballast is driven using critical discontinuous mode (CDCM) operation of both the input and output stages. In CDCM, the current through the input inductor L1 is switched on and off so that it ramps up and down creating a triangular type wave form during each high frequency switching cycle. The envelope of the triangle peaks outlines the desired waveform, and the actual waveform produced, after filtering, is the desired sinusoidal waveform. The output stage is also operated in CDCM such that the resulting current through inductor L2 also has a high frequency triangular waveform. The envelope of the triangle peaks outlines the desired waveform, and the actual waveform produced after filtering is the desired square waveform.
By utilizing CDCM and a constant charging time, the amplitude of the input current directly follows a prescribed portion of the supply voltage. It is only necessary to measure the zero crossing of the input inductor current so that the system can maintain the triangular waveform. The prior art feedback loop which continuously monitors the input current is eliminated and a simple zero crossing sensor is utilized instead. In an enhanced embodiment, the PWM switches are switched on and off in a manner such that lossless switching occurs by timing the switching correctly.
In a further enhanced embodiment, a saturable transformer is added in order to monitor when the inductor current reaches zero and properly time the switching. The complexity required to monitor zero crossings in an inductor current is significantly less than that of continuously monitoring the value of an inductor current and feeding it back for an adjustment, therefore simplifying the complexity, and reducing the cost, of the circuitry.
Similarly, by utilizing CDCM, the amplitude of the output current directly follows the DC bus voltage with polarity alternating with the mains voltage as controlled by the switching of switches M3 and M4. Again, it is only necessary to measure the zero crossings of the inductor current, thus eliminating the feedback loop. With proper switch timing, lossless switching can be achieved.
Further advantage and functionalities of the present invention will become apparent from review of the following detailed description and drawings of an exemplary embodiment of the present invention.